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TM
MP6001
Monolithic Flyback/Forward DC-DC Converter
The Future of Analog IC Technology
TM
INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
DESCRIPTION
The MP6001 is a monolithic Flyback/Forward DC-DC converter which includes a 150V power switch and is capable of delivering up to 15W output power. It can also be used for boost and SEPIC applications. The MP6001 uses the fixed-frequency peak current mode primary controller architecture. It has an internal soft-start, auto-retry, and incorporates over current, short circuit, and over-voltage protection. The MP6001 can also skip cycles to maintain zero load regulation. It has a direct optocoupler interface which bypasses the internal error amplifier when an isolated output is desired. The MP6001 is ideal for telecom applications, and is available in a compact, thermally enhanced SO8 package with an exposed pad.
FEATURES
* * * * * * * * * * * * * * * * Integrated 0.9 150V Power Switch Cycle-by-Cycle Current Limiting Programmable Switching Frequency Duty Cycle Limiting with Line Feed Forward Integrated 100V Startup Circuit Internal Slope Compensation Disable Function Built-in Soft-Start Line Under Voltage Lockout Line Over Voltage Protection Auto-Restart for Opened/Shorted Output Zero Load Regulation Thermal Shutdown Telecom Equipment VoIP Phones, Power over Ethernet (PoE) Distributed Power Conversion
APPLICATIONS
"MPS" and "The Future of Analog IC Technology" are Trademarks of Monolithic Power Systems, Inc.
TYPICAL APPLICATION
+VIN 36V~72V D1 1N4148 D2 B330A
5V @ 3A
VOUT
Efficiency vs Load Current
90 80
VIN = 36V
EFFICIENCY (%)
70 60 50 40 30 20
PC357
2 6 3 4 LINE VCC SW VIN 8 7 1 5
VIN = 48V VIN = 75V
MP6001
FB COMP GND RT
C3 10nF
R3
TL431 -VIN
MP6001_TAC_S01
10 0 0 0.5 1.0 1.5
VOUT = 5V
2.0 2.5 3.0
LOAD CURRENT (A)
MP6001-EC01
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
PACKAGE REFERENCE
TOP VIEW
GND LINE FB COMP 1 2 3 4 8 7 6 5 SW VIN VCC RT
ABSOLUTE MAXIMUM RATINGS (1)
VSW............................................. -0.5V to +180V VIN ............................................. -0.3V to +120V All Other Pins.............................. -0.3V to +6.5V Junction Temperature...............................150C Lead Temperature ....................................260C Storage Temperature ..............-65C to +150C
Recommended Operating Conditions
(2)
MP6001_PD01_SOIC8N
Supply Voltage VCC ........................... 4.5 V to 6V Output Voltage VSW.................... -0.5V to +150V Input Voltage VIN ........................ -0.3V to +100V Operating Temperature .............-40C to +85C
Part Number* MP6001DN *
Package SOIC8N
Temperature -40C to +85C
Thermal Resistance
(3)
SOIC8N (Exposed Pad) ......... 50 ...... 10... C/W
Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1" square of 1 oz copper.
JA
JC
For Tape & Reel, add suffix -Z (eg. MP6001DN-Z) For RoHS compliant packaging, add suffix -LF (eg. MP6001DN-LF-Z)
ELECTRICAL CHARACTERISTICS
VCC = 5.0V, VLINE = 1.8V, RT = 10k, TA = +25C, unless otherwise noted.
Parameter Quiescent Supply Current Line OV Threshold Voltage Line OV Hysteresis Line UV Threshold Voltage Line UV Hysteresis VCC Upper Threshold Voltage VCC Lower Threshold Voltage Feedback Voltage Feedback Input Current Error Amplifier Gain Bandwidth (4) Error Amplifier DC Gain (4) Comp Output Source Current Comp Output Sink Current Switch-On Resistance Switch Leakage Current Minimum Oscillating Frequency Maximum Oscillating Frequency Thermal Shutdown (4) Thermal Shutdown Hysteresis (4) Current Limit (4) Startup Current Symbol Condition ICC 1.2V < VLINE < 3.2V, VFB = 1.3V VCC = 5.0V VCC = 5.0V VCC = 5.0V VCC = 5.0V Min 2.85 1.16 5.75 4.30 1.16 VFB = 1.2V 1 60 VFB = 1.0V, VCOMP = 0.5V VFB = 1.4V, VCOMP = 2.5V VSW = 0.1V VSW = 150V RT = 100k RT = 10k 2 2 0.9 1 55 550 150 30 2 3 Typ 1.0 3 300 1.21 100 6.0 4.50 1.21 50 Max 1.5 3.15 1.26 6.25 4.70 1.26 Units mA V mV V mV V V V nA MHz dB mA mA A KHz KHz C C A mA
VFB IFB GBW AV IOH IOL RON ILK FMIN FMAX
ILIM Ist
VIN = 20V, VCC = 4.0V
Note: 4) Guaranteed by design, not production tested.
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
PIN FUNCTIONS
Pin # 1 2 3 4 5 6 7 8 Name Description Ground. Power return and reference node. UV/OV Set Point. Short to ground to turn the controller off. Regulation Feedback Input. Inverting input of the error amplifier. The non-inverting is internally FB connected to 1.2V COMP Error Amplifier Output. Oscillator Resistor and Synchronous Clock Pin. Connect an external resistor to GND for RT oscillator frequency setting. It can be used as a synchronous input from external oscillator clock. VCC Supply Bias Voltage. VIN High Voltage Startup Circuit Supply. Output Switching Node. High voltage power N-Channel MOSFET drain output. The internal SW start bias current is supplied from this pin. GND LINE
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 48V, VOUT = 5V, TA = +25C, unless otherwise noted.
Steady State Test
VIN = 48V, VOUT = 5V, IOUT = 2.7A
Synchronize Programmable Oscillator
fSW = 54KHz, 60KHz of SYNC signal is applied to RT pin.
Synchronize Programmable Oscillator
fSW = 54KHz, 500KHz of SYNC signal is applied to RT pin.
VOUT 2V/div.
VOUT AC Coupled 50mV/div.
VOUT 2V/div. VSW 20V/div.
VSW 100V/div.
IL 1A/div.
VSYNC 2V/div. ITRANS 1A/div.
VSW 20V/div. VSYNC 2V/div. ITRANS 1A/div.
MP6001-TPC01
MP6001-TPC02
MP6001-TPC03
Duty Cycle vs Line Voltage
70
Short Circuit Test
Over current hiccup at VIN = 48V, IOUT = 4.4A
MAXIMUM DUTY CYCLE (%)
60 50 40 30 20
VOUT AC Coupled 50mV/div.
VSW 100V/div. VCC 2V/div.
ILOAD 1A/div.
VOUT 2V/div. IL 1A/div.
1.0
1.5
2.0
2.5
3.0
MP6001-TPC05
100ms/div.
MP6001-TPC06
LINE VOLTAGE (V)
MP6001-TPC04
Start-up through Enable
VIN = 48V, VOUT = 5V, IOUT = 2.7 Resistive Load
VOUT 1V/div. VCC 2V/div. VSW 50V/div. ITRANS 1A/div.
Shut-down through Enable
VIN = 48V, VOUT = 5V, IOUT = 2.7A Resistive Load
VCC 2V/div. VSW 50V/div. VOUT 1V/div. ITRANS 1A/div.
4ms/div.
MP6001-TPC07
4ms/div.
MP6001-TPC08
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
OPERATION
The MP6001 uses programmable fixedfrequency, peak current-mode PWM with a single-ended primary architecture to regulate the output voltage. The MP6001 incorporates features such as protection circuitry and an integrated high voltage power switch into a small 8-pin SOIC. This product targets high performance, cost effective DC-DC converter applications.
6 VCC + -LINE 2 3.0V + -OVLO REGULATOR IBIAS REF UVLO 7 VIN STARTUP 6.5V 4.5V
1.2V
+ --
COMP 4 ERROR AMPLIFIER 1.2V FB 3 + -EA
THERMAL MONITOR CONTROL LOGIC
8 SW
-PWM COMPARATOR +
1 GND
SOFT-START CURRENT LIMIT CLOCK RT 5 OSC SLOPE COMP + 1.0V --
-+ CURRENT LIMIT COMPARATOR LEB CURRENT SENSE
MP6001_BD01
Figure 1--Functional Block Diagram
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE High Voltage Startup The MP6001 features a 100V startup circuit, see Figure 1. When power is applied, the capacitor at the VCC pin is charged through the VIN pin. When the voltage at the VCC pin crosses 6.0V without fault, the controller is enabled. The VCC pin is then disconnected from the VIN pin and VCC voltage is discharged via the operating current. When VCC drops to 4.5V, the VCC pin is reconnected to the VIN pin and VCC will be recharged. The voltage at the VCC pin repeats this ramp cycle between 4.5V and 6.0V. The VCC pin can be powered with a voltage higher than 4.5V from an auxiliary winding to reduce the power dissipated in the internal startup circuit. The VCC pin is internally clamped at 8V. Under-Voltage and Over-Voltage Detection The MP6001 includes a line monitor circuit. Two external resistors form a voltage divider from the input voltage to GND; its tap connects to the LINE pin. The controller is operational when the voltage at the UV/OV pin is between 1.2V and 3V. When the voltage at the UV/OV pin goes out of this operating range, the controller is disabled and goes into standby mode. The LINE pin can also be used as a remote enable. Grounding the UV/OV pin will disable the controller. Error Amplifier The MP6001 includes an error amplifier with its non-inverting input connected to internal 1.2V reference voltage. The regulated voltage is fed back through a resistor network or an optocoupler to the FB pin. Figure 2 shows some common error amplifier configurations.
6 VCC D1 C1 R1 FB 3 PRIMARY WINDING R3 R2 C2 C3 4 1.2V +
EA
--
COMP
(a) Using Primary winding to provide feedback
6 VCC 1.2V C2 FB 3 R3 R2 4 + -EA
COMP
(b) Feedback is from Secondary (Common Collector)
6 VCC 1.2V C2 FB 3 4 + -EA
COMP
(c) Feedback is from Secondary (Common Emitter)
MP6001_F02
Figure 2--Error Amplifier Configurations
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE Synchronize Programmable Oscillator The MP6001 oscillating frequency is set by an external resistor from the RT pin to ground. The value of RT can be calculated from:
RT = 10k x 550KHz fS
The MP6001 can be synchronized to an external clock pulse. The frequency of the clock pulse must be higher than the internal oscillator frequency. The clock pulse width should be within 50ns to 150ns. The external clock can be coupled to the RT pin with a 100pF capacitor and a peak level greater than 3.5V. Duty Cycle Limiting with Line Feed Forward The MP6001 has a DMAX (maximum duty cycle) limit at 67.5% when the LINE pin voltage is equal to 1.3V. As VLINE increases, DMAX reduces. Maximum duty cycle can be calculated by:
2 .7 V D MAX = x 100% 2.7 V + VLINE
Auto-Restart When VCC is biased from an auxiliary winding and an open loop condition occurs, the voltage at the VCC pin increases to 6.5V. When VCC crosses the threshold voltage, the auto-restart circuit turns off the power switch and puts the controller in standby mode. When VCC drops to 4.5V, the startup switch turns on to charge VCC up again. When VCC crosses 6.0V, the switch turns off and the standby current discharges VCC back to 4.5V. After repeating the ramp cycles between the two threshold voltages 15 times, the auto-restart circuit is disabled and the controller begins soft-start.
Limiting the duty cycle at high line voltage protects against magnetic saturation and minimizes the output sensitivity to line transients.
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
APPLICATION INFORMATION
Switching Frequency The frequency (fS), has big effects on the selection of the transformer (Tr), the output cap, (C2), and the input cap, (C1). The higher the frequency, the smaller the sizes for Tr, C2, and C1. However, a higher frequency also leads to higher AC power losses in the power switch, control circuitry, transformer, and in the external interconnection. The general rule states that lower the output power, higher the optimum switching frequency. For low current (<10A) applications, fS is usually 200KHz to 300KHz if synchronous rectifiers are used and 300KHz to 500KHz if Schottky rectifiers are used. Fundamental Equations The transformer turns ratio N is defined as:
N= NP NS
Where NP and NS are the number of turns of the primary and secondary side windings, respectively. The output voltage VO is estimated to be:
VO = V D x IN 1- D N
Transformer (Coupled Inductor) Design 1. Transformer Turns Ratio The transformer turns ratio determines the duty cycle range, selection of the rectifier (D2), primary side peak current, primary snubber loss, and the current as well as voltage stresses on the power switch (S). It also has effects on the selection of C1 and C2. A higher transformer turns ratio (N) means the following: * Higher Duty Cycle * Higher voltage stress on S (VDS), but lower voltage stress on D2 (VD2). * Lower primary side RMS current (IS(RMS)), but higher secondary side RMS current (ID2(RMS)). * Use of a smaller input capacitor but bigger output capacitor. * Lower primary side peak current (IS(PEAK)) and lower primary snubber loss. * Lower main switch (S) turn-on loss For a 5V power supply design, with VIN=36V~75V, below table shows the voltage stresses of the power switch (S) and the rectifier (D2). Table 1--Main Switch (S) and Rectifier (D2) Voltage Stress vs. Transformer Turns Ratio
N 4 5 6 7 8 9 10 11 DMAX 0.36 0.41 0.45 0.49 0.53 0.56 0.58 0.60 VDS (V) 119 125 131 138 144 150 156 163 VDS/0.9 (V) 132 139 146 153 160 167 174 181 VD2 (V) 38 32 28 25 23 21 20 19 VD2/0.9 (V) 42 36 31 28 26 24 22 21
Where D is the duty cycle. The steady-state drain to source voltage of the primary power switch when it is off is estimated as:
VDS = VIN + N x VO
The steady-state reverse voltage Schottky diode D2 is estimated as:
VD2 = VO + VIN N
of
the
The output current is calculated as:
IO = ID x (1 - D)
Note: The voltage spike due to the leakage inductance of the transformer and device's voltage rating/derating factors were considered. See power switch selection and snubber design for more information.
Where ID is the average current through Schottky diode when it is conducting. The input current is calculated as:
IIN = IS x D
Where IS is the average current through the primary power switch when it is conducting.
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE 2. Ripple Factor of the Magnetizing Current The conduction loss in S, D2, the transformer, the snubber, and in the ESR of the input/output capacitors will increase as the ripple of the magnetizing current increases. The ripple factor (Kr) is defined as the ratio of the peak-to-peak ripple current vs. the average current as shown in Figure 3.
I Kr = M IM
Solid wire, Litz wire, PCB winding, Flex PCB winding or any combination thereof can be used as transformer winding. For low current applications, solid wire is the most cost effective choice. Consider using several wires in parallel and interleaving the winding structure for better performance of the transformer. The number of primary turns can be determined by:
NP = L F x IP B MAX x A E
Where IM can be derived either from input or output current;
I I0 IM = IN = D N x (1 - D)
ID2/N IM
Where BMAX is the allowed maximum flux density (usually below 300mT) and AE is the effective area of the core. The air gap can be estimated by:
Gap = o x N2 x A E LF
IM
0
DTS
TS
MP6001_F03
Figure 3--Magnetic Current of Flyback Transformer (Reflected to Primary Side) The input/output ripple voltage will also increase with a high ripple factor, which makes the filter bigger and more expensive. On the other hand, it can help to minimize the turn-on loss of S and reverse-recovery loss due to D2. With nominal input voltage, Kr can be selected at 60%~120% for most DC-DC converters. The primary side (or magnetizing) inductance can be determined by:
LF = VIN x D x TS K r x IM
5. Right Half Plane Zero A Flyback converter operating in continuous mode has a right half plane (RHP) zero. In the frequency domain, this RHP zero adds not only a phase lag to the control characteristics but also increases the gain of the circuit. Typical rule of thumb states that the highest usable loop crossover frequency is limited to one third the value of the RHP zero. The expression for the location of the RHP zero in a continuous mode flyback is given by:
fRHPZ = R LOAD x (1 - D) 2 x N2 2 x L F x D
Where RLOAD is the load resistance, LF is the magnetizing inductance on transformer primary side, and N is the transformer's turn ratio. Reducing the primary inductance increases the RHP zero frequency which results in higher crossover frequencies.
3. Core Selection Pick a core based on experience or through a catalog (Refer to http://www.ferroxcube.com). Select an ER, EQ, PQ, or RM core to minimize the transformer's leakage inductance. 4. Winding Selection
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE Duty Cycle Range The duty cycle range is determined once N is selected. In general, the optimum operating duty cycle should be smaller for high input/low output than low input/high output applications. Except for high output voltage or wide input range applications, the maximum D usually does not exceed 60%. Voltage Stress of the Internal Power Switch & External Schottky Diode For the internal power switch, the voltage stress is given by: VDS = VIN + VO x N + VP Where VP is a function of LLK (leakage inductrance), fS, R, C, CDS, VIN, IO, etc. Please refer to Figure 4. The lower the LLK and Io, the lower the Vp. Smaller R can reduce Vp, but power loss will increase. See Snubber Design for details. Typically VP can be selected as 20~40% of (VIN+NVO).
LLK VC C1 S -- C + R Tr D2 C2 D IS + VDS -ID2
can be selected (VO+VIN/N), thus:
VPD2
as
40~100%
of
VDS(MAX) = K s x ( VIN(MAX) + NV0 )
Where KS=1.2~1.4, and
VD 2(MAX ) = K D 2 ( V0 + VIN(MAX ) N )
Where KD2=1.4~2. For example,
VIN(MAX ) = 75 V, N = 8, K S = 1.25, K D2 = 1.6, VO = 5 V VDS = 1.25 x (75 V + 8 x 5 V ) = 144 V VD2 = 1.6 x (5 V + 75 V / 8) = 23 V
So
the power switch rating should be higher than 144V, and the rated voltage for the synchronous rectifier or Schottky diode should be higher than 23V. Snubber Design (Passive) Snubber for Power Switch Figure 5 shows four different ways to clamp the voltage on the power device. RCD type of snubber circuit is widely used in many applications.
S
RD CD
S
DZ
(A)
(B)
VC VDS
VP
CD RD DZ
VIN 0
MP6001_F04
S
S
(C)
(D)
MP6001_F05
Figure 4--Key Operation Waveform For the rectifier, D2, the voltage stress is given by:
VD2 = VO + VIN + VPD 2 N
Figure 5--Snubber Designs
Use of a R-C or R-C-D type snubber circuit for D2 is recommended.
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE RCD Type of Snubber Design Procedure: 1. Setting VP Higher VP means higher voltage stress on the power switch, but lower power loss. Usually, VP can be set as 20%~40% of (VIN+ NxVO).
VP VC N x VO VIN VDS 0
MP6001_F06
For a given AC ripple voltage, VIN_PP, C1 can be derived from:
C1 = IIN x (1 - D) x TS VIN _ PP
VIN_PP may affect the C1 voltage rating and converter stability. C1 RMS current has to be considered:
IRMS _ C1 = IIN x (1 - D) D
C1 has to have enough RMS current rating. Output Filter The simplest filter is an output capacitor (C2), whose capacitance is determined by the output ripple requirement. The current waveform in the output capacitor is mostly in rectangular shape. The full load current is drawn from the capacitors during the primary switch on time. The worse case for the output ripple occurs under low line and full load conditions. The ripple voltage can be estimated by:
V0 -PP -C = IO x D C2 x f S
Figure 6--Voltage Waveform of Primary Power Switch Shown in Figure 5(C) 2. Estimated RCD snubber loss is given by:
PRCD _ LOSS = PLK x (1 + N x VO ) VP
Where:
PLK = 1 2 L LK x IP x f C 2
PLK is the energy stored in the leakage inductance (LLK), which carries the peak current at the power switch turn-off. 3. Calculate values of the RD and CD of RCD snubber by:
RD = VP
2
PRCD _ LOSS
1 >> fS
ESR also needs to be specified for the output capacitors. This is due to the step change in D2 current results in a ripple voltage that is proportional to the ESR. Assuming that the D2 current waveform is in rectangular shape, the ESR requirement is then obtained by given the output ripple voltage.
VO -PP _ RESR = IO x ESR (1 - D)
R D x CD
The total ripple voltage can be estimated by:
VO -PP = VO -PP -C + VO -PP _ ESR
Input Capacitor The input capacitors (C1) are chosen based upon the AC voltage ripple on the input capacitors, RMS current ratings, and voltage rating of the input capacitors.
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE Control Design Generally, telecom power supplies require the galvanic isolation between a relatively high input voltage and low output voltages. The most widely used devices to transfer signals across the isolation boundary are pulse transformers and optocouplers.
D Tr RESR CO VO RLOAD
The MP6001 uses current mode control to achieve easy compensation and fast transient response. A type II compensation network which has two poles and one zero is needed to stabilize the system. The practical compensation parameters are provided in the EV6001DN datasheet. Boost Controller Application The MP6001 can be used as a boost controller as shown in Figure 8.
VIN D1 200V/1A 180V 20mA
1 8 7
VIN
+ -S
d
VCC
R5 R1
GND
SW
2
LINE
VIN
R6 + --+ VREF R4 R3
MP6001
R2 C1
3 FB VCC 6
4
COMP
RT
5
TL431
Rb
C3 10nF
MP6001_F07
Figure 7--Simplified Circuit of Isolated Power Supply with Optocoupler Feedback
MP6001 _F08
Figure 8--High Voltage LED Boost Controller Circuit
MP6001 Rev. 0.91 4/5/2006
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MP6001 - MONOLITHIC FLYBACK/FORWARD DC-DC CONVERTER INITIAL RELEASE - SPECIFICATIONS SUBJECT TO CHANGE
PACKAGE INFORMATION
SOIC8N
PIN 1 IDENT. 0.229(5.820) 0.244(6.200)
0.150(3.810) 0.157(4.000)
0.0075(0.191) 0.0098(0.249)
SEE DETAIL "A"
NOTE 2 0.013(0.330) 0.020(0.508) 0.050(1.270)BSC 0.011(0.280) x 45o 0.020(0.508)
0.189(4.800) 0.197(5.004) 0.053(1.350) 0.068(1.730) 0.049(1.250) 0.060(1.524)
0o-8o
0.016(0.410) 0.050(1.270)
DETAIL "A"
SEATING PLANE 0.001(0.030) 0.004(0.101) NOTE: 1) Control dimension is in inches. Dimension in bracket is millimeters. 2) Exposed Pad Option Only (N-Package) ; 2.55+/- 0.25mm x 3.38 +/- 0.44mm. Recommended Solder Board Area: 2.80mm x 3.82mm = 10.7mm2 (16.6mil2)
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.
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